A steep increase in communication traffic with spreading use of new services such as moving image distribution and the like utilizing cloud computing and Internet is expected. Studies and development of an optical transmitter-receiver that may transmit a signal which is as fast as 100 Gbps are now being carried forward in order to cope with the communication traffic which is being increased.
However, if the bit rate per one wavelength is increased, the signal quality will be degraded owing to a reduction in proof stress to optical signal to noise ratio (OSNR) and waveform distortion caused by wavelength dispersion, polarization mode dispersion or nonlinear effect of a transmission path. Therefore, attention is now being paid to an optical digital coherent receiving system that has proof stresses to the OSNR and wavelength distortion of the transmission path.
Since the optical digital coherent receiving system allows improvement of proof stress to the OSNR, compensation for waveform distortion using a digital signal processing circuit and adaptive equalization to a temporal variation of the propagation characteristic of an optical transmission path, characteristics of high quality may be obtained even in transmission performed at a high bit rate.
In addition, the optical digital coherent receiving system is a system that extracts light intensity and phase information by a coherent receiving system and quantizes the extracted light intensity and phase information using an ADC (Analog-Digital Converter) to demodulate them using a digital signal processing circuit unlike an existing system of allocating ON-state light intensity and OFF-state light intensity to a binary signal to perform direct detection.
DP-QPSK (Dual Polarization-Quadrature Phase Shift Keying) which is one of phase modulation systems used in the optical digital coherent receiving system allows to allocate two-bit data to four modulated optical phases (0 degrees, 90 degrees, 180 degrees and 270 degrees) for each of P-polarized light and S-polarized light. Since in the DP-QPSK, the symbol rate may be reduced to one-fourth the information transfer rate, downsizing and cost saving of the system may become possible.
FIG. 1 is a block diagram illustrating an example of a configuration of an optical digital coherent receiver.
A light signal that has been input through an optical fiber 10 is split into a P-polarized light signal and an S-polarized light signal which are orthogonal to each other using a polarization beam splitter (PBS) 11. The polarized light signals so split using the polarization beam splitter 11 are respectively input into 90° optical hybrids 13 and 14. The optical digital coherent receiver also includes a locally oscillating light source 15 that locally oscillates light of a predetermined frequency. The light oscillated from the locally oscillating light source 15 is split into the P-polarized light and the S-polarized light using the polarization beam splitter 12. The polarized lights so split using the polarization beam splitter 12 are respectively input into the 90° optical hybrids 13 and 14. Here, the light signals respectively input into the 90° optical hybrids 13 and 14 mach the light oscillated from the locally oscillating light source in polarization. That is, when the light signal which is input into the 90° optical hybrid 13 from the polarization beam splitter 11 is the P-polarized light, the light output from the polarization beam splitter 12 is also the P-polarized light. Likewise, since the light signal which is input into the 90° optical hybrid 14 from the polarization beam splitter 11 is the S-polarized light in the above mentioned case, the light from the polarization beam splitter 12 is also the S-polarized light.
The light signal which has been sent through the optical fiber 10 and then split and polarized is mixed with the light which has been oscillated from the local light emitting source 15 and then split and polarized in the 90° optical hybrids 13 and 14. A phase modulation component in the light signal is converted to a change in intensity of the light signal by mixing the light signal with the light oscillated from the locally oscillating light source. A photoelectric converter 16 converts the light signals from the 90° optical hybrids 13 and 14 to electric signals. The received signals so converted to the electric signals using the photoelectric converter 16 are converted to digital signals using an ADC unit 17. The digital signals so converted using the ADC unit 17 are digitally subjected to signal processing using a digital signal processing circuit 25.
In the digital signal processing circuit 25, first, a process of removing signal distortion from the signal sent from the ADC unit 17 is digitally performed using a waveform distortion compensator 18. The waveform distortion compensator 18 removes a fixed component of distortion based on the characteristics of the optical fiber 10 in the signal distortion that the light signal has been suffered owing to wavelength dispersion, polarization mode dispersion, nonlinear effect and the like while it is being propagated through the optical fiber 10.
An output from the waveform distortion compensator 18 is input into a phase adjustor 19. In the phase adjustor 19, the timing of sampling the received signal is adjusted. If the sampling timing of the received signal matches the symbol transition timing, it will become difficult to correctly detect the signal value. Therefore, the sampling timing is adjusted to avoid such a situation as mentioned above. The phase adjustor 19 receives a result of timing detection from a timing regenerator 20 that detects the sampling timing and generates a timing adjustment signal. The timing adjustment signal is sent to a frequency variable oscillator 21 that generates a clock signal to be sent to the ADC unit 17. The frequency variable oscillator 21 adjusts the frequency and the phase of the clock signal on the basis of the timing adjustment signal and inputs the clock signal into the ADC unit 17. The ADC unit 17 samples the received signal on the basis of the clock signal so adjusted in frequency and phase.
An output from the phase adjustor 19 is input into an adaptive equalizer 22. The adaptive equalizer 22 performs a process of removing the signal distortion caused by aged deterioration or the like of the optical fiber 10 on the signal. The signal distortion removing process that the waveform distortion compensator 18 has performed is a fixed compensating process which is not sufficient to completely remove signal distortion components. Therefore, the adaptive equalizer 22 compensates for the signal distortion components which are left uncompensated.
An output from the adaptive equalizer 22 is input into a light source frequency offset estimation and compensation unit 23. The light source frequency offset estimation and compensation unit 23 performs a process of stopping rotation of a signal point on an I-Q plane caused by an offset of the frequency of the light from the locally oscillating light source 15 from the frequency of the received signal sent through the optical fiber 10. This process is performed in order to eliminate a fixed phase deviation of a complex signal included in the output signal from the adaptive equalizer 22.
An output from the light source frequency offset estimation and compensation unit 23 is input into a carrier phase offset estimation and compensation unit 24. The light source frequency offset estimation and compensation unit 23 has performed the process of removing a difference between the frequency of the received signal and the frequency of the locally oscillated light. Although rotation of the signal point on the I-Q plane is stopped by performing the above process, the signal point may possibly stay at a 90-deg-rotated position simply by performing the above mentioned process. Therefore, the carrier phase offset estimation and compensation unit 24 performs a process of removing a phase offset occurred owing to 90-deg-rotation of the signal point on the I-Q plane on the input signal.
The light source frequency offset estimation and compensation unit 23 is a circuit that detects a light source frequency offset which will be desired to remove a light frequency deviation (offset) occurred between a signal light source of a transmitter and a locally oscillated light source of a receiver and includes a loop filter and the like.
In an adaptive equalization type filter which is the kind of the loop filter, if much noise is included in an input signal into that filter, it will sometimes occur that a convergent value of a filter coefficient to be updated approaches zero (0) and hence it becomes difficult to obtain a signal which is accurate enough to be used as an output from the filter as in the case of the loop filter included in the light source frequency offset estimation and compensation unit 23.
FIG. 2 is a diagram illustrating an example of a light source frequency offset estimation circuit according to related part.
In the example in FIG. 2, all processes to be performed using preceding units of an arg ( ) unit 38 are performed by complex number (I, Q) arithmetic operations and the arg ( ) unit 38 arithmetically operates the phase (the argument) of a complex signal output from its last preceding unit.
The loop filter used for estimation of a frequency offset amount is an oblivion average type IIR filter having a coefficient (1−α)<<1.0.
The light source frequency offset estimation and compensation unit 23 includes a light source frequency offset estimation circuit 29 illustrated in FIG. 2 and a unit that removes the frequency offset using a frequency offset value which is output from the light source frequency offset estimation circuit 29. The complex signal which is output from the adaptive equalizer 22 is input into the light source frequency offset estimation circuit 29. The complex signal is a signal that includes an I signal as a real number component and a Q signal as an imaginary number component and when the term “complex signal” is used, it means a set of an I signal and a Q signal in reality. The input complex signal is input into an intersymbol phase difference detection unit 32. The input complex signal is also input into a one-symbol delayer 31 and then input onto the intersymbol phase difference detection unit 32 after delayed by one symbol using the delayer. The intersymbol phase difference detection unit 32 arithmetically operates a difference between the two input complex signals. The difference signal is obtained by performing complex subtraction and is in the form of a complex signal that includes the phase difference between two symbols in its phase component. A signal obtained by subtracting an output from a loop filter 30 from this intersymbol phase difference signal is output from the intersymbol phase difference detection unit 32.
The output from the intersymbol phase difference detection unit 32 is input into the loop filter 30. In the loop filter 30, the complex signal is multiplied by (1−α) using a multiplier 34 and is input into an adder 35. Here, α is a previously determined value and a real number value that satisfies a relation 1>α>0. The adder 35 performs complex addition and adds a value obtained by multiplying an output from a phase rotation amount vector storage unit 36 by α (the same as the above mentioned coefficient) using a multiplier 37 to an output from the multiplier 34.
The phase rotation amount vector storage unit 36 is configured using a register and stores an output from the adder 35. The adder 35 calculates a formula (an output from the intersymbol phase difference detection unit)×(1−α)+(data in the phase rotation amount vector storage unit)×α to arithmetically operate an average of signals input into the loop filter 30. That is, although in general, a weighted average of x and y is obtained from (a×x+b×y)/(a+b) wherein a and b are weights, when a+b=1, a=1−b and hence the above formula for the average is obtained. Thus, the average value obtained using the adder 35 is a weighted average value. Thus, the output from the intersymbol phase difference detection unit and (1−α) are multiplied using the multiplier 34, the data in the phase rotation amount vector storage unit and α are multiplied using the multiplier 37, and the obtained multiplied values are added together to obtain the average value which is weighted by the above a and b. The output from the phase rotation amount vector storage unit 36 is an average of the signals input into the loop filter 30 and an angle component (a phase component) of the average signal (the phase rotation amount vector) on an I-Q plane is arithmetically operated using the arg ( ) unit 38 and is output as a frequency offset value. The frequency offset value is used to remove the frequency offset included in the received signal.
An output from the loop filter 30 is fed-back to the intersymbol phase difference detection unit 32 and is used to be subtracted from an intersymbol phase difference obtained using the intersymbol phase difference detection unit 32. In the example, since the frequency offset is removed using a succeeding unit of the arg ( ) unit 38 using the output from the loop filter 30, subtraction of the phase difference corresponding to the above operation is performed by subtracting the output from the loop filter 30 from the intersymbol phase difference.
Various frequency offset detection and estimation circuits are known in the related art.